原创 EMI in Switched Mode Power Supplies (SMPS)

EMI in Switched Mode Power Supplies (SMPS)<?xml:namespace prefix = o ns = "urn:schemas-microsoft-com:office:office" />

开关电源的电磁干扰

Switched Mode Power Supplies are usually a part of a complex electronic system. The system operates with electric signals with much lower amplitude and energy compared to those on an SMPS. It means that usually the SMPS is the strongest electrical noise generator in the whole system.

开关电源通常会是一部电子设备的组成部分。电子设备里运行的电信号相对开关电源里的在幅值和能量上要小得多。这意味着开关电源通常是电子设备里的最强干扰源。

Especially the power switches with their high dv/dt and di/dt switching slopes are the sources of EMI. The source of differential mode interferences is the current switched by a MOSFET or a diode. High rates of dv/dt and parasitic capacitors to the ground are the reasons for common mode interferences.

尤其功率开关对电压和电流的高速切换是电磁干扰的源头。差模干扰来源于场效应晶体管或二极管对电流的开关。高速率的电压变化和对地的分布电容是产生共模干扰的原因。

Factors of influence on EMI spectrum

电磁干扰频谱的影响要素

Different parameter can affect the EMI spectrum of an SMPS, e.g. the switching slopes of the power semiconductors, the operating point regarding switch voltages and currents. Some of these influence factors will be analyzed in this chapter.

不同的因素都能影响开关电源的电磁干扰频谱，举例来说，功率半导体的开关速度，运行点的开关电压和电流。此章将分析这些干扰因素。

Test setup

测试装置

The results presented in this chapter have been made using the shown test setup (Fig. 9), if not other specified. The test setup is a chopper without an EMI filter using 380mΩ/<?xml:namespace prefix = st1 ns = "urn:schemas-microsoft-com:office:smarttags" />11A/600V CoolMOS and Silicon Carbide Schottky diode. It can be operated in Discontinuous and Continuous Conduction Modes.

除非另外说明，此章呈现的结果来自图９所展示的测试装置。这个测试装置是使用３８０毫欧姆内阻、１１安培漏极电流、６００伏特漏源极耐压的CoolMOS和碳化硅肖特基二极管组成的无电磁干扰滤波器的断路器。它可以工作在断续和连续电感电流模式。

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Fig. 9 Test chopper

图９　测试断路器

External gate resistance

外置的栅极驱动电阻

The well-known way to control the switching speed and therefore the EMI behavior is a variation of the external gate resistance. If the resistance will be increased, the time constant determined by this resistance and the capacitance of the MOSFET will be increased as well. The switching transient will be slowed down and thereby the electrical noise becomes lower. Fig. 10 shows the controllability of the di/dt and dv/dt values for turn on and turn off transients of CoolMOS SPP11N60C3.

众所周知，控制开关速度来改变电磁干扰的方法是改变栅极驱动电阻。如果电阻增加，由这个电阻和场效应晶体管的栅极电容决定的时间常数将增加。开关过程变慢导致电磁噪声下降。图１０展示了SPP11N60C3型号的CoolMOS在不同栅极电阻驱动下开通和关断过程的电流、电压变化速率。

典型漏极电流变化速率

随栅极驱动电阻变化的漏极电流变化速率，感性负载，１２５摄氏度结温，３８０伏特漏源极电压，０至正１３伏特栅源极电压，１１安培的漏极电流

典型漏源极电压变化速率

随栅极驱动电阻变化的漏源极电压变化速率，感性负载，１２５摄氏度结温，３８０伏特漏源极电压，０至正１３伏特栅源极电压，１１安培的漏极电流

Fig. 10 Typical switching slopes versed gate resistance of CoolMOS SPP11N60C3

图１０　典型随CoolMOS SPP11N60C3栅极驱动电阻变化的漏极开关变化速率

As it can be seen the drain current and drain to source voltage slopes can be adjusted simply using the external gate resistor during turn on and off transients. The noise emission can be also controlled in this way in high frequency range (Fig. 11 and Fig. 12).

显而易见，开通和关断过程的漏极电流、漏源极电压变化速率可以简单的通过外置栅极驱动电阻进行调整。在高频范围的噪声辐射也可通过此方法控制（图１１和图１２）。

Fig. 11 Influence of the external gate resistance (discontinuous current mode, Id=6A, measured)

图１１　外置栅极驱动电阻的影响（断续电感电流模式，６安培漏极电流）

Fig. 12 Influence of the external gate resistance (continuous current mode, Id=6A, measured)

图１2　外置栅极驱动电阻的影响（连续电感电流模式，６安培漏极电流）

Gate-source and drain-source voltage

栅源极电压和漏源极电压

The appropriate spectrum for 6 A peak drain-current, 13 V gate voltage and 380 V drain-source voltage has been chosen as reference.

为在峰值６安培漏极电流下获得适当的频谱，选择１３伏特的栅源极电压和３８０伏特的漏源极电压为参考。

Different maximum values of the gate-source voltage (or Vcc of the control IC) were investigated – 10V, 13V and 15V.

不同的栅源极电压最大值（或是控制集成电路的电源电压）被选择——１０伏特，１３伏特和１５伏特。

Also the variation of the bus voltage from 380V to 200V has been measured.

同样从３８０伏特到２００伏特的直流母线电压被选定。

Fig. 13 shows that a change of the gate voltage from 10V to 15V hardly influences the spectrum. It can be explained by the fact, that 10V is still high enough to fully open the MOSFETs channel. Further reduction to values close to Miller plateau for the given operating point will slow down the switching and reduce the spectra.

图１３展示出栅源极电压从１０伏特到１５伏特变化几乎不对干扰频谱产生影响。他可以解释这样的事实：１０伏特已经足够开通金属氧化物场效应晶体管的导电沟道。进一步减小米勒电场将导致开关速率变慢并降低干扰频谱幅值。

Fig. 13 Influence of the gate and drain-source voltage (discontinuous conduction mode, Id=6A, Rgate="6".8 Ohm, measured)

图１３　栅源极电压和漏源极电压的影响（断续电感电流模式，６安培漏极电流，６.８欧姆栅极电阻）

The decrease of the drain-source voltage or bus voltage affects the entire spectrum evenly according to Fourier theory. It means that the spectrum will decrease with the drain-source voltage.

降低漏源极直流母线电压影响干扰信号按傅立叶展开式的全部频带。这意味着干扰频谱幅值随漏源极电压的降低而降低。

Drain current

漏极电流

The peak values of the drain current have been varied in discontinuous (2A, 6A, 11A) and continuous (2A, 6A) conduction modes.

漏极的峰值电流被设定为断续电感电流模式下的２安培、６安培、１１安培和连续电感电流模式下的２安培。

Only little differences in the frequency range under 1 MHz at different current peak values (Fig. 14 and Fig. 15) can be noticed except from some resonance frequencies. The drain current was not decreased enough to operate the CoolMOS switch in the linear field of its output characteristic. Thus the dv/dt of the switch remained the same and thereby the spectrum changes can be hardly recognized.

除了几个谐振频率，在１兆赫兹以下的频带，干扰信号在不同峰值电流下测试出来的结果仅有微小的差别（图１４和图１５）。CoolMOS开关的漏极电流没有减小到足够进入输出特性的线性区域。因此相同的电压变化速率几乎不对干扰信号的频谱产生影响。

Fig. 14 Influence of the drain current (discontinuous current mode, Rgate="6".8 Ohm, measured)

图１４　漏极电流的影响（断续电感电流模式，６.８欧姆栅极驱动电阻）

Fig. 15 Influence of the drain current (continuous conduction mode, Rgate="6".8 Ohm, measured)

图１５漏极电流的影响（连续电感电流模式，６.８欧姆栅极驱动电阻）

Flyback converter example

反激变换器的例子

Analysis of basic waveforms

基本波形分析

The analysis of the basic waveforms will be done on a simulated example of a flyback converter operating in discontinuous conduction mode. Typical drain-source voltage waveform of the primary side switch is shown in Fig. 16.

在电感电流断续模式下运行的反激变换器的典型一次侧漏源极开关电压波形见图１６。

Fig. 16 Typical drain-source voltage of the MOSFET in a flyback

图１６　反激变换器的典型漏源极电压

These drain-source voltage waveforms can be theoretically distinguished into typical elements. Different physical phenomena influence the waveform at given time interval. Fig. 17 and Tab. 4 demonstrate the main elements of the voltage waveform. The superposition of all these elements results in a typical drain-source voltage shown in Fig. 16.

这些漏源极电压波形能用典型的理论来描述。各个时间段有不同物理现象影响这些波形。图１７和平台４描述了电压波形的主要原理。把这些原理按时序整合呈现出图１６所示的典型漏源极电压。

Fig. 17 Main elements of the drain-source voltage

图１７　漏源极电压的主要原理

原理１：开通期间的电压下降过程

原理２：在开通期间因寄生震荡产生的电流尖刺

原理３：关断期间的电压上升

原理４：缓冲电路的钳位电压

原理５：钳位过程结束后主要由场效应晶体管输出电容和变压器漏感引起的寄生振荡

原理６：磁芯存储磁能释放完毕后主要由场效应晶体管输出电容和变压器电感引起的寄生振荡

原理７：反激变换器释放磁能期间的反射电压

原理８：与直流母线电压等幅的主要方波

Tab. 4 Main elements of the drain-source voltage

平台４　漏源极电压的主要原理

The spectrum of the whole drain-source waveform (Fig. 16) is presented in Fig. 18.

图１６所示的漏源极电压呈现的电磁干扰频谱见图１８。

Fig. 18 Spectrum of the drain-source voltage (as shown in Fig. 16)

图１８　图１６所示的漏源极电压呈现的电磁干扰频谱

The spectra of the main elements of the drain-source voltage can be found in Fig. 20. Fig. 19 is exactly the same as Fig. 17 and has been repeated here for better under-standing.

图２０描述了漏源极电压主要原理产生的电磁干扰频谱。为便于理解，将图１７映射成图１９。

Fig. 19 Main elements of the drain-source voltage (repeated, same as Fig. 17)

图１９　漏源极电压的主要原理（正确重复　图１７）

Fig. 20 Spectra of the main elements of the drain-source voltage

图２０　漏源极电压主要原理产生的电磁干扰频谱

This method allows associating certain parts of the spectrum with their root causes, i.e. the peak at 20 MHz in the spectrum of the drain-source voltage is caused by the parasitic oscillation due to the output capacitance of the MOSFET and the leakage inductance of the transformer.

这种方法可以确定电磁干扰频谱中某些频点的来源，也就是说漏源极电压产生的电磁干扰频谱中的２０兆赫兹峰点是钳位过程结束后主要由场效应晶体管输出电容和变压器漏感引起的寄生振荡产生的。

The analysis of the drain current of the primary switch will be done in the same way. Fig. 21 demonstrates a typical drain current in a DCM flyback.

对一次侧开关的漏极电流进行分析采用相同的方法。图２１展示出一个工作于电感电流断续模式反激变换器的典型漏极电流。

Fig. 21 Typical drain current in a flyback

图２１　反激变换器的典型漏极电流

This waveform can be presented as a superposition of the following elements (Fig. 22 and Tab. 5). The superposition of all these elements results in a typical drain current shown in Fig. 21.

这个波形可以被看作是下列原理的叠加（图２２和平台５）。全部这些波形的叠加整合结果变成图２１所示的典型漏极电流。

Fig. 22 Main elements of the drain current

图２２　漏极电流的主要原理

原理１：漏极电流的主要三角波形

原理２：在开关开通期间因寄生分布电容引起的电流尖刺

原理３：钳位过程结束后主要由场效应晶体管输出电容和变压器漏感引起的寄生振荡

原理４：磁芯存储磁能释放完毕后主要由场效应晶体管输出电容和变压器电感引起的寄生振荡

Tab. 5 Main elements of the drain current

平台５　漏极电流的主要原理

The spectrum of the whole drain current waveform (Fig. 21) is presented in Fig. 23.

全部漏极电流波形产生的电磁干扰频谱（图２１）呈现在图２３。

Fig. 23 Spectrum of the drain current (as shown in Fig. 22)

图２３　漏极电流产生的电磁干扰频谱（与图２２相同）

The spectra of the main elements of the drain current can be found in Fig. 25. Fig. 24 is exactly the same as Fig. 22 and has been repeated for better understanding.

漏极电流主要原理产生的电磁干扰频谱见图２５。图２４和图２２相同。

Fig. 24 Main elements of the drain current

图２４　漏极电流的主要原理

Fig. 25 Spectra of the main elements of the drain current

图２５　漏极电流主要原理产生的电磁干扰频谱

As in case of drain-source voltage this method allows to associate the elements of the drain current waveform with its contribution to the whole spectrum. For example, the peak at 20 MHz in the spectrum is caused by the parasitic oscillation due to the output capacitance of the MOSFET and the leakage inductance of the transformer.

就象漏源极电压的例子那样，用这种方法也可以找出漏极电流的哪一部分对电磁干扰频谱产生影响。举例说明，２０兆赫兹的峰点是钳位过程结束后主要由场效应晶体管输出电容和变压器漏感引起的寄生振荡产生的。

This method of separating the waveform in time domain into its main elements helps to find out what part of the spectrum in frequency domain caused by what related physical phenomena. The separation into main elements should be done in respect of reasonable events in the power circuit like on and off slopes, oscillations, clamping, snubbering, reflected voltage, etc.

这种在时域里对主要原理进行拆分的方法有助于找出产生电磁干扰频段的干扰源。这种离析主要原理的手法有助于合理审视电源电路里诸如变化速率、振荡、钳位、缓冲、反射电压等过程。

In this flyback example only the primary switch has been analyzed as active source of electrical noise. There are also others, like secondary side diodes or synchronous rectifier, control IC (especially its gate drive), etc. In order to obtain more complete analysis all these interference sources have to be analyzed.

在这个反激变换器里只对一次侧开关进行电磁噪声产生的分析。但是还有其他的部分，象二次侧的二极管或同步整流器、控制集成电路（尤其是它们的栅极驱动）等等。按顺序分析将获得更完善的关于这些电磁干扰源的解析。

However, it is impossible to predict the conducted EMI spectrum using this approach due to the fact, that only interference sources are considered. There is no analysis of the spreading paths of the interference in this method.

然而，这种方法不可能预知用频谱反映的电磁干扰的实际行为，仅仅是干扰源被重视起来。在那里没有对分布参数产生的干扰进行分析的方法。

Nevertheless, the association of harmonics root cause with the respected physical phenomena will reduce the efforts of EMI reduction. The impact of the identified root cause can be reduced not only by filtering, but also by means of influencing the root cause itself.

不过，重视物理现象并不能成就电磁干扰的降低。降低干扰并不仅仅是滤波，也同样意味着干扰源自身的影响。

Operation modes of discontinuous flyback converter

电感电流断续工作反激式变换器的运行模式

　　The flyback converter running in discontinuous conduction mode can be operated in hard switching or quasi resonant (or valley switching, or ZVS) mode regarding the primary side switch. The difference between a hard switching and quasi resonant flyback converter is the turn on time point of the primary switch. In a hard switching mode the turning on of the MOSFET is not synchronized with the drain-source voltage value. This type of converters runs mainly in fixed frequency mode.

电感电流断续工作的反激式变换器一次侧开关可工作于硬开关或准谐振（或谷值开关或零电压开关）模式。硬开关和准谐振反激变换器之间的差异在于一次侧开关的开启时间点。在硬开关里场效应晶体管的开启波形拐点并不和漏源极电压值同步。这种变换器大体上运行于固定频率模式。

In a quasi resonant mode the resonant circuit determined by the output capacity of the MOSFET and the inductance of the transformer will be utilized to switch on at lowest possible value of the drain-source voltage. This circuit starts to oscillate at the end of the current flow through the secondary side of the transformer, hence at the end of the flyback phase. The MOSFET will be turned on at the minimum of this oscillation. The quasi resonant approach uses this oscillation to achieve minimum voltage switching during turn on for the MOSFET. This operation mode runs at a variable frequency.

在准谐振模式里，由变压器电感和场效应晶体管输出电容引起的谐振促使开关的开通时刻发生在漏源极电压的最小值上。这种电路在电流从变压器二次侧流尽以后（反激回扫过程结束）开始振荡。场效应晶体管将在振荡幅值的最小值开启（谷值开通）。这种运行模式工作在可变的频率上。

Higher amplitude of the oscillation results in lower drain source voltage level at which the MOSFET turns on correspondingly lower switching losses and higher efficiency of the system.

更高幅值的振荡导致场效应晶体管更低的漏源极开通电压幅值来产生更低的开关损耗和更高的系统效率。

To achieve high oscillation peaks, the design of the transformer has to be set to high reflected voltage. This increase of the reflected voltage results in a higher drain-source voltage blocking MOSFET and longer duty cycles.

要达到比较高的振荡电压峰值，变压器的反射电压必须设置的比较高。增加的反射电压导致使用更高漏源极击穿电压的场效应晶体管和更大的开关占空比。

Comparison of three different flyback solutions has been made. All of them have been operation at 300 kHz, bus voltage of 400 V, output power of 120 W, output voltage of 16 V. These design included different modes of operation and different values of reflected voltage, resulting in different MOSFET’s voltage ratings:

比较现有的三种反激变换器。它们都工作在３００千赫兹，直流母线电压４００伏特，输出功率１２０瓦特，输出电压１６伏特。这些设计包含不同的运行模式和反射电压等级，因此使用不同电压等级的场效应晶体管：

l      Hard switching flyback with CoolMOS 600V, reflected voltage of 100V

l      硬开关反激变换器使用６００伏特CoolMOS，１００伏特反射电压

l      Quasi resonant flyback with CoolMOS 600V, reflected voltage of 100V

l      准谐振反激变换器使用６００伏特CoolMOS，１００伏特反射电压

l      Quasi resonant flyback with CoolMOS 800V, reflected voltage of 390V

l      准谐振反激变换器使用8００伏特CoolMOS，39０伏特反射电压

The clamping snubber circuit was set to the rated breakdown voltage of the MOSFET (600 V and 800 V respectively).

钳位缓冲电路被设定在场效应晶体管的额定击穿电压上（分别为６００伏特和８００伏特）。

Flyback in hard switching mode with 600V MOSFET

使用６００伏特场效应晶体管的硬开关反激变换器

The hard switching approach (as shown in Fig. 26) doesn’t consider the minimum drain-source voltage. The MOSFET will be turned on hard, in this case at a voltage level of 500 V (at time point 3.3 μs). The discharge of circuits’ parasitic capacitances leads to a high current spike during turning on.

硬开关（图２６所示）几乎不考虑漏源极电压的最小值。场效应晶体管开通应力大，在这个例子里，开通电压在５００伏特（在３.３微秒的时间点）。由寄生电容引起的泄放电流在开通时产生很高的电流尖刺。

Fig. 26 Drain-source voltage and drain current of hard switching 600V flyback

图２６　６００伏特硬开关反激变换器的漏源极电压和漏极电流

Flyback in quasi resonant mode with 600 V MOSFET

使用６００伏特场效应晶体管的准谐振反激变换器

The drain-source voltage (Fig. 27) starts oscillating at the end of the flyback phase and reaching the minimum of 300 V when the MOSFET turns on.

漏源极电压（图２７）在反射过程结束后并减小到３００伏特时场效应晶体管导通。

The duty cycle is lower compared to an 800 V solution due to a lower reflected voltage of 100V. Shorter duty cycle for the same output power results in higher peak currents on the primary side.

因为１００伏特的反射电压，比较８００伏特解决方案它有更小的占空比。小占空比实现同样的功率输出必须使用更高的一次侧峰值电流。

Fig. 27 Drain-source voltage and drain current of quasi resonant 600V flyback

图２７　６００伏特准谐振反激变换器的漏源极电压和漏极电流

Flyback in quasi resonant mode with 800 V MOSFET

使用８００伏特场效应晶体管的准谐振反激变换器

The drain-source voltage (Fig. 28) starts oscillating at the end of the flyback phase and reaching the minimum of 100V when the MOSFET turns on. The turning on current spike is low.

漏源极电压（图２８）在反射过程结束后并减小到１００伏特时场效应晶体管导通。开通电流尖刺比较低。

The duty cycle is higher compared to a 600V solution due to a higher reflected voltage of 390V. Longer duty cycle for the same output power results in lower peak currents on the primary side.

因为有３９０伏特的反射电压，所以有比６００伏特解决方案更大的占空比。更大的占空比实现同样的输出功率可以使用更低的一次侧峰值电流。

Fig. 28 Drain-source voltage and drain current of quasi resonant 800V flyback

图２８　８００伏特准谐振反激变换器的漏源极电压和漏极电流

Comparison of spectra

干扰频谱比较

The spectra of the drain-source voltages for corresponding flyback design (Fig. 26Fig. 27 and Fig. 28) are shown in Fig. 29.

相应设计的反激变换器（图２６、图２７和图２８）的漏源极电压干扰频谱如图２９所示。

Fig. 29 Spectra of the drain-source voltage (simulated)

图２９　漏源极电压的频谱（仿真）

As it can be seen the voltage spectrum of the 800V quasi resonant flyback is higher at frequencies below 1 MHz, and is getting lower above 1 MHz compared to both 600V designs. This can be explained by two major differences of the 800V drain-source voltage waveform. First, the clamping voltage during the MOSFETs turning off is 800V, what is higher then of 600V. It leads to higher harmonics amplitudes in lower frequency range. Second, the turn on occurs in voltage minimum due to quasi resonant switching, which results in lower spectrum in higher frequency range.

和６００伏特设计比较，８００伏特准谐振反激变换器的电压频谱在１兆赫兹以下更高一点，在１兆赫兹以上开始变小。这里有两条理由可以解释８００伏特漏源极电压波形的两个差异。第一，在场效应晶体管关断期间钳位电压是８００伏特，高于６００伏特。它产生低频段的高振幅。第二，开通发生在准谐振开关电压的最小值，这导致更高频段频谱中更低的幅值。

Due to the fact, that the 800V quasi resonant flyback has lower peak current, its spectrum is significantly lower across almost complete frequency range.

事实上，８００伏特准谐振反激变换器拥有更低的峰值电流，它的频谱意味着在全频带有更低的幅值。

The 800V quasi resonant design with lower current peak and lower drain-source voltage during turning on of the MOSFET demonstrates advantages in conducted EMI spectra regarding the primary side.

拥有更低峰值电流和场效应晶体管漏源极开通电压的８００伏特准谐振设计展示出一次侧传导电磁干扰降低的优势。